Adaptive load sharing system

ABSTRACT

Sharing of load current in desired ratios between multiple electrical voltage sources is achieved without coupling between the sources by controlling a selection switch to select each voltage source for a proportion of the time at a high rate, the switching rate components being prevented from being seen by either the load or the sources through use of low-pass filters.

BACKGROUND

The present invention relates to load sharing between two electricenergy sources, such as a DC to AC inverter on the one hand and theutility grid on the other hand, and in particular, such as a DC to ACconverter deriving its DC energy input from solar panels or asolar-charged battery

In the prior art, two types of DC to AC converter for use in solarenergy installations are known.

One type of inverter, called a grid-tie inverter, converts DC energy toan AC current and feeds the current back into the electricity grid toreduce net energy consumption. Such an inverter is said to operate in“current mode” as the voltage is defined by the utility grid to which itis connected, and it is controlled to output a current at that voltage,the product of which is the desired power level to be transferred fromthe DC source to the grid.

The big disadvantage of this method is that, if the utility is in anoutage, the system must be switched off. Another disadvantage is that,as the penetration of solar energy increases, there will come a pointwhere the grid cannot accept all the aggregated power that solar systemsare attempting to feed back into it on a sunny day. Other disadvantagesinclude political issues such as the requirement for connectionpermission from the utility, permitting from local or State authorities,utility subversion of solar via unattractive billing regimes andbarriers to innovation such as the National Electrical Code and Listingrequirements.

A second type of inverter, called a standalone inverter, converts DCpower to an AC voltage equivalent to the grid voltage which can be usedto power loads directly, thereby avoiding consuming power from the grid.Energy storage, i.e. a battery, is required for standalone systems tosmooth out the different between supply and demand as clouds move overthe sun uncorrelated with appliance loads being switched on and off.

U.S. Pat. No. 8,937,822 to Applicant Dent, entitled “Solar EnergyConversion and Utilization System”, discloses both grid-tie andstandalone DC to AC inverters, and discloses an adaptively controlleddistribution panel that selects which breaker circuits to power from astandalone inverter and which to power from the electric utility grid,thereby assisting in dynamically matching supply and demand.

When it is attempted to utilize a standalone inverter to power a home ina totally off-grid installation, a very large and expensive battery isrequired to ensure the ability to bridge a period of overcast weather.However, the above '822 patent discloses a system that uses both gridand solar power and adaptively transfers load between utility and solarin such a way that the net solar power used in place of utility powermatches the average solar power received over a shorter period of time,thus allowing the battery size and cost to be drastically reduced.

Another method of using a current-mode inverter that is known in theprior art is called “utility assist mode”. This is substantially thesame as grid-tie mode, except that the inverter current is controlled tosupplement current coming from the grid without ever feeding power backto the grid. The current is thus controlled to be less or equal to thecurrent consumed from the grid by active appliances. The advantage ofthis method is that a finer degree of matching between solar powergenerated and solar power used can be achieved. However, since theinverter is still in parallel with the grid, it must switch off if thegrid is in outage, unless a fast disconnect switch operates andsimultaneously switches to voltage-mode DC to AC inversion. For thelatter, a battery will also be required for energy smoothing. It is alsonot clear that connection permission is not required, as the invertercurrent waveform is being impressed on the grid and must meet utilityspecifications and all of the requirements of specification UL1741.There is therefore a need for an alternative method of ensuring the useof just as much solar power as is momentarily available while avoidingever connecting the inverter to the grid. The requirement may besummarized in general as a method and apparatus for continuously sharingload between two energy sources without a direct electrical connectionbetween the two energy sources.

SUMMARY

A first AC electrical voltage source such as the electric utility gridis connected through a first low-pass filter to a first input of anelectronic two-way AC selection switch. A second AC electrical voltagesource such as a solar-energy-driven DC to AC inverter is connectedthrough a second low-pass filter to a second input of the two-way ACselection switch. The second AC voltage source is roughly synchronizedin phase and voltage with the first voltage source. The selection switchis controlled by a controller to select one of the first or secondvoltage sources to be output from the switch output terminal. The switchoutput terminal is connected to one or more electrical loads through oneor more optional third low pass filters. The switch controller operatesthe switch to select between the first and second voltage sourcesalternately with a controlled mark-space ratio and at a frequency higherthan the-cutoff frequency of the low-pass filters such that the meancurrents drawn respectively from the first and second voltage sourcesare sinusoidal at the power frequency and in relative proportion to thecontrolled mark-space ratio. The sharing of load current between thefirst and second voltage sources may thus be adaptively controlled basedon data provided to the switch controller. The data may for examplerelate to the amount of solar energy available to power the DC to ACinverter.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a prior art “Utility Assist” inverter arrangement.

FIG. 2 shows a utility-assist system with a standalone capability

FIG. 3 shows the inventive utility-assist system with a standalonecapability

FIG. 4 shows the inventive load sharing circuit for a single phase

FIG. 5 shows the effect of rapidly selecting between roughlysynchronized sources

FIG. 6 shows an equivalent transformer circuit

FIGS. 7A, B and C illustrate phasor diagrams for combining sourcevoltages

FIG. 8 shows the target operating region and the disallowed region

FIG. 9 shows a fast, semiconductor AC power switch suitable for theinvention

FIG. 10 shows more details of the switches and low pass filters.

FIG. 11 shows addition of Spike Catching diodes.

FIG. 12 shows a transistor switching sequence that avoids switchingspikes.

FIG. 13 shows a microprocessor-based timing controller circuit.

FIG. 14 shows another possible AC switch circuit

FIG. 15 shows an electromechanical method of combining power sources

DETAILED DESCRIPTION

FIG. 1 shows a prior art connection for a utility assist inverterarrangement. Only live (hot) wiring is shown and neutral and groundwiring is omitted for simplicity, but shall of course be providedaccording to electrical code requirements.

An input from an electric utility grid is received through electricmeter (100) to main service entrance panel (200). In the USA,residential service is 120/240 volt AC 60 Hz split-phase, having two180-degree out-of-phase hotlegs of 120 volts with 240 volts availablebetween them.

The service entrance panel contains service disconnect breakers and loadbreakers, and may be the only panel in some installations. FIG. 1however shows an installation having also a sub-panel (300), which isfed from the service entrance panel through 100 amp feeder breakers. Inthe exemplary installation of FIG. 1, some loads are fed directly fromthe service panel and excluded from the loads which may be partly fed bythe solar inverter. The remainder of the loads that may be partly fedfrom the solar inverter are fed from sub-panel (300) and the output fromDC-to-AC inverter (400) is fed into a 40 amp back-feed breaker insub-panel (300). The total current taken by sub-panel (300) from theservice entrance panel (200) is measured by current measuring device(401) and is equal to the total load current supplied from sub-panel(300) minus the load current supplied from inverter (400).

Now, if the load current supplied from sub-panel (300) is partly on onephase and partly on the opposite phase, while the current supplied byinverter (400) is on a single phase, it may only be allowed to back-offthe current taken from the same utility phase as that to which theinverter is connected. In fact a single-phase inverter is capable ofbacking-off current taken from both utility phase by relying on theutility pole transformer to transfer current to the other phase;however, the current taken from one phase could then be negative whilethe other is positive, and this may not be permitted or may not behandled correctly by meter (100), depending on its design. If a 240-voltDC-to-AC inverter is connected through a two-pole breaker in sub-panel(300), the same issue can arise when the load on one phase is less thanthe load on the opposite phase. Thus for either of the above-describedinverter connections, utility validation and acceptance of theinstallation would be needed in the form of “connection permission”. Insome States, such a connection permission may entail a higher standingmonthly charge to be paid to the utility company, which could negate theeconomics of installing this type of solar system

In order to back-off utility consumption on both phases by current fromDC-to-AC inverter (400), it would be necessary to use an inverter havingtwo independently controllable anti-phase outputs. This wouldeffectively be two inverters. Each output would be controlled tominimize the utility current drawn on the phase that it was supplying,as measured by current measuring device (401) while preventing thatcurrent from ever becoming negative. If the current is permitted tobecome negative, the installation is no longer merely a utility assistsystem, but is a grid-tied system, which may require a different type ofconnection permission and a different billing regime.

When DC-to-AC inverter (400) is fed from solar panels (500) without anenergy storage device such as a battery, it cannot be guaranteed to beable to supply any particular number of loads as the amount solarinsolation varies substantially and is of course zero at night.Therefore such an installation as in FIG. 1 must shut down if theutility fails. If it desired to continue to power loads when the utilityis in outage, several modifications to FIG. 1 are required. The mostimportant modification is that the connection between the utility andthe powered loads must be broken. This can be done by inserting anautomatically controlled service disconnect switch in the feeder cablefrom the feeder breakers in service entrance panel (200) to sub-panel(300).

The second modification is to insert an energy storage device (e.g. abattery) between the solar panels (500) and inverter (400). The thirdchange is that the inverter must be of a type known as bimodal, whichoperates as a current source when in utility assist mode and as avoltage source when operating to power loads alone.

FIG. 2 shows another utility-assist configuration that is capable ofpowering loads during a utility outage. Storage battery (600) has beenadded between the DC output of solar panels (500) and the DC input ofInverter (400) in order to supply load power in the event of a utilityoutage. Several devices or functions that may customarily or optionallybe included in the DC side of such an installation, including groundfault detection, arc fault detection, Maximum Power Point Tracking andbattery charge controllers are not shown for simplicity, but may beadded if required. These functions are disclosed in more detail in U.S.Pat. Nos. 9,190,836 and 8,937,822 to Applicant, which are herebyincorporated by reference herein in their entirety.

A second significant change between the prior art of FIG. 1 and thesystem of FIG. 2 is the introduction of isolation switch (402) todisconnect the utility supply if the utility should fail. Moreover,current measuring device (401) and isolation switch (402) have beenrelocated in a more logical position in the system, namely into inverter(400), in order to permit sub-panel (300) to be a standard sub-panel.

Sub-panel 300 now has a clear function: Since Inverter (400) will nowattempt to power some but not all loads during a utility outage, theloads that it can support are fed from sub-panel (300) while the loadsit cannot support are fed from service entrance panel (200). When theloads fed from sub-panel (300) are a particularly limited subset of allloads, sub-panel (300) is sometimes called “a subsistence panel”. Whenthe inverter has more power than a bare “subsistence” load sub-set wouldrequire however, the term “subsistence panel” may not adequatelydescribe the extent of power that continues to be available during autility power outage.

Among the current measurements that current measuring device (401) mayperform, the important ones are the net currents flowing throughisolation switch (402) and current measuring device (401) betweenservice entrance panel (200) and the parallel connection of inverter(400) and sub-panel (300). It will be appreciated that within currentmeasuring device (401), like phases (i.e. L1 or L2) from each of serviceentrance panel (200), inverter (400) and sub-panel (300) are effectivelyconnected together in a Y junction. Nevertheless, the current in any orall of the three branches of each Y junction may be measured for eachphase L1 and L2, making six potential current measurements in total. Themeasurement of net current flowing from the utility in each phase isused to control the inverter such that the net current drawn from theutility through the depicted 100A sub-panel feeder breakers is minimizedbut not permitted to become negative (i.e. not permitted to represent areverse power flow to the grid). When however the grid fails, inverter(400) may supply however much current is required to power the loads insub-panel (300).

Correct operation of the system of FIG. 2 is fraught with a number ofdifficulties as follows:

(a) Current taken by loads varies as appliances are switched on and off.Therefore when an appliance is switched off, the inverter maymomentarily be causing reverse power flow to the grid during the time ittakes to adapt down to the new load current. Thus prevention of reversepower flow can not be perfectly achieved. This may require the system tohave the same connection permissions, unfavorable billing regime andonerous UL listing requirements as a grid-tied system.

(b) Detection of failure of the utility is required in order to throwisolation switch (402) and direct inverter (400) to supply loads with acontrolled voltage. If the utility grid fails, because it is connectedto the output of inverter (400), other upstream installations on thesame utility supply branch are likely to so overload the inverter thatits output voltage will collapse. This is indeed one way to detect thatthe utility has failed, but it is not instantaneous, and does notprevent the loads seeing at least a brown-out before action is taken andisolation switch (402) is thrown. If many similar installations exist inthe same neighborhood, the phenomenon of “Islanding” can occur and thenit may not be possible to detect a utility failure by a voltagebrown-out alone. Instead, a frequency measurement may be performed andIslanding is indicated by a gradual drift of the AC frequency outsidepreset limits.

(c) Isolation switch 402 must handle the total possible feeder currentof 100 A and is therefore likely to be a large and mechanically slowcontactor. The duration of the brown-out may be sufficient to causecomputers to reboot and alarm clocks to blink.

It is the purpose of the current invention to provide an improvedresponse to utility failure while achieving all the advantages of autility-assist system and avoiding the above disadvantages. This isachieved by a novel circuit that allows two electric power sources toshare the burden of driving common loads without ever being connectedtogether.

FIG. 3 shows the inventive utility-assist system with standalonecapability. The differences from FIG. 2 to FIG. 3 are that the isolationswitch (402) has been removed and the current measuring device (401),which joined like phases of the inverter, sub-panel (300) and utilitypower from service entrance panel (200) in a Y-connection has beenreplaced by unit (450) which connects utility power from serviceentrance panel (200) and inverter (400) to sub-panel (300) in a new andinventive way. The inventive connection allows 60 Hz AC power to flowfrom inverter (400) and from service entrance panel (200) to the loadsin sub-panel (300) simultaneously without ever making a connectionbetween the utility feed from service entrance panel (200) and the poweroutput connections of inverter (400). This feat may sound seeminglyimpossible until having examined the operation of the inventive,adaptive load-sharing device (1000) which is explained in more detailwith the aid of FIG. 4.

FIG. 4 illustrates an inventive load-sharing device (1000) having afirst low pass filter (1001) receiving an input form phase L1 of a firstelectrical power source, a second low pass filter (1002) having an inputfor the same phase L1 of a second electrical power source, and a thirdlow pass filter (1003) for delivering a combined L1 output to a load.

Low pass filters are two-port devices generally having a two-terminalinput port and a two-terminal output port. The input and output portsmay share a common terminal, e.g. ground, which, however, in the case ofdevice (1000) is the neutral connection associated with the powersources and loads. The input port of filter (1001) is the L1 power andneutral from the first source; the input port of filter (1002) is the L1power line and neutral of the second source, and the output port offilter (1003) is the output L1 line and neutral to the loads.

The output port of filter (1001) passes through AC switch (1010-1) tothe input port of filter (1003) and the output port of filter (1002)passes through AC switch (1010-2) to the input port of filter (1003). Aslong as both AC switches are never closed at the same time, there istherefore never a connection between the L1's of the first and secondpower sources. Switch timing controller (1002) controls the closing ofswitches (1010-1) and (1010-2) while imposing the condition that theyshall never be closed at the same time. In fact, controller (1020) cancontrol switch (1010-1) to open substantially at the same time as itcontrols switch (1010-2) to close and vice versa, thus ensuring that oneor other of the first or second power source is always connected throughlow pass filter (1003) to the load.

AC switches (1010-1, 1010-2) can be fabricated with fast-switchingsemiconductor devices, as will be further explained below. Timingcontroller (1020) can operate such switches to close alternately at veryhigh frequencies, for example 100,000 times per second or more, and in aspecified mark/space ratio defined by an external control signal appliedto timing controller (1020). For example, if the switches are operatedalternately at a cycle frequency of 100 KHz, that is a periodicity of 10uS, switch (1010-1) may be closed for 3 uS out of the 10 uS period whileswitch (1010-2) is open, alternating with switch (1010-2) being closedfor 7 uS while switch (1010-1) is open. Thus current is drawn from thefirst source 3/10ths of the time while current is drawn form the secondsource 7/10ths of the time. When low pass filters (1001,1002) havecut-off frequencies substantially lower than the 100 KHz exemplary cyclefrequency, the power sources simply see a mean current demand of 0.3 and0.7 of the load respectively. Thus by controlling the mark/space ratio,it is possible to control the proportion of the load current borne bythe first and second power sources.

FIG. 5 illustrates the case where the AC voltage waveforms of the firstand second power sources are only approximately synchronized. The firstand second sources supply sinusoidal voltage waveforms which, asillustrated, are of approximately the same voltage amplitude butslightly displaced in phase. Switches (1010-1,1010-2) select rapidlybetween the two sources producing the tessellated waveform bounded bythe two source waveforms. If the two waveforms were accuratelysynchronized, the tessellations would represent a negligible voltageexcursion when switching from one source to the other source, and no lowpass filter (1003) would be needed to remove the tessellations. The useof low pass filter (1003) however removes the high-frequencytessellations even when the sources are only approximately synchronized,producing a smoothed sinusoidal voltage into the load which is simply

(aV1+bV2)/(a+b) where a/b is the ratio of the time for which source 1 isselected to the time for which source 2 is selected.

Assuming now a load impedance of Z=(R+jX) ohms, the load current wouldbe

Iload=(aV1/Z+bV2/Z)/(a+b)=(aI1+bI2)/(a+b) where I1 is the current thatwould flow if source 1 were selected all the time and I2 is the currentthat would flow if source 2 were selected all the time. The actualcurrent is split between the two sources in the same way as the meanload voltage is split between V1 and V2. This suggests that thearrangement is electrically equivalent to the transformer seriescoupling of the two sources to the load as shown in FIG. 6.

FIG. 6 shows a transformer (703) having primary-to-secondary turns ratioof (a+b):a to reduce the voltage from source 1 (701) from V1 toaV1/(a+b). A similar transformer (704) with turns ratio b:(a+b) reducesthe voltage from source 2 (702) from V2 to bV2/(a+b). The twosecondaries are connected in series to produce a voltage (aV1+bV2)/(a+b)into load 705 causing a current I to flow. When the two voltages V1 andV2 are both equal to V, it may be seen that the voltage applied to theload is also equal to V. It may also be seen that if a=b, then thevoltage delivered to the load is the mean of V1 and V2.

The transformers also divide the load current so that a current ofaI/(a+b) flows from source 1 and a current bI/(a+b) flows from source 2.The main difference in character between the circuits FIG. 5 and FIG. 6however is that in FIG. 5, the turns ratios are continuously variable bychanging the mark/space ratio of switch selection.

We now examine with the aid of FIG. 6 the circumstances under whichpower could flow backwards into one source from the other.

FIGS. 7 A,B and C illustrate with a phasor diagrams the combination ofthe voltages from two power sources by the transformer equivalentcircuit of FIG. 6, and by equivalence, also by the circuit of FIG. 5.

FIG. 7A illustrates the case where the two source voltages V1 and V2 areroughly equal, but the factor “a” is 55% and the factor “b” is 45%. V1is taken as the reference phase (0 degrees) and so is horizontal vector.If V2 was in phase with V1, the resultant Vload would be 100% of V1 orV2, which is the desired operating point. When V2 has a substantialphase difference compared to V1 however, the resultant Vload is lessthan 100% and has a phase angle between that of V1 and V2, asillustrated in FIG. 7A. If the load is resistive, the angle of Vload isalso the angle of the load current. It is seen that this has a positiveprojection on V1, implying that power is drawn from the source V1. Ifall possible phase angles of V2 are explored, the resultant can lieanywhere on the circle of FIG. 7A, but the resultant always has apositive projection on V1, and so power from source V1 is alwayspositive.

In FIG. 7B, the scaling factors “a” and “b” are such the scaled valuesof V1 and V2 are equal. The circle traced out by all possible phaseangles of V2 relative to V1 now passes through the origin, at whichpoint the load voltage would be zero and no current would be drawn fromV1 or V2.

In FIG. 7B, the scaled value of V2 is greater than the scaled value ofV1. Now the circle traced out by exploring all phases of V2 crosses intothe left half plane, shown shaded, which is the area where the resultanthas a negative projection on V1 implying power is being fed to thesource V1 in this area.

Applying the conclusions of FIG. 7 to the problem in hand, it may beconcluded that power can only be fed back to the grid from the solarinverter (400) of FIG. 3 if its output voltage exceeds that of the gridand it is grossly mis-synchronized.

Ensuring that the inverter voltage is nearly equal to or less than thegrid voltage and that it is roughly in phase synchronism with the gridare two conditions that are much easier to monitor and control thanensuring that the prior art current-mode inverter (400) of FIG. 1 neverfeeds power back to the grid. Moreover, the Inverter may now operate involtage-mode all the time, which is the mode required to power loadswhen the utility is in outage.

FIG. 8 summarizes the regions of relative voltage and relative phaseconstituting the desired operating region, shown as shaded region A, andthe disallowed region of reverse power flow, annotated as Region B.Since it is desired to be able to power the load either entirely fromthe utility (V1) or entirely from the solar inverter (V2), V1 and V2should be within +/−5% of each other to avoid any noticeable change inappliance operation. Furthermore, when powered half from utility andhalf from solar (a=b=50%), the combined voltage should also be within 5%of normal. This occurs with a phase mis-synchronisation between V1 andV2 of as much as +/−36 degrees when V1=V2, thus giving rise to shadedregion A as the target operating region. When powering half from utilityand half from inverter with V1=V2 and a=b=50%, this corresponds to theV1/V2=1 circle of FIG. 8, showing that no reverse power flow can occurunless the phase error were the maximum 180 degrees, and this would notsatisfy the load voltage regulation requirements either.

When powering ⅔rds from solar and ⅓rd from utility with V1=V2, the ratioof the combined voltages is on the V1/V2=0.5 circle of FIG. 8. The phaseerror must still be a very large 120 degrees before reverse power flowfrom inverter to utility occurs. Even powering nearly 100% from invertercauses no utility reverse power flow with phase errors as much as 90degrees.

One advantage of the invention is that the voltage V1 of the utility andthe voltage V2 of the inverter can be measured totally independently ofeach other, as the one does not affect the other, unlike the prior artof FIG. 2 where utility and inverter are connected directly in parallel.Thus the relative voltage and relative phase of utility and inverter areeasy to measure independently and continuously when using the invention.This also allows instantaneous detection of a utility brown-out and asubstantially instantaneous reaction to fill the brown-out with inverterpower, thus achieving much cleaner power.

FIG. 9 shows one arrangement for power switches (1010-1 and 2 of FIG. 4)suitable for use in the invention. Two MOSFETs (1100,1200) are connectedback-to-back at their source terminals (s). Their drain terminals (d)become the switch contacts. Since the circuit is symmetrical, it doesn'tmatter which drain terminal is considered the input and which theoutput. The gates (g) are also connected together and the connectedgates are driven with respect to the connected sources by a gate driverchip (1300) powered from an isolated power supply of the order of 10-15volts DC.

Gate driver chips generally provide isolation between the control logicinput and the output, allowing the control on/off logic signal to comefrom a control circuit that is not isolated, and may be connected toground.

The operation of the circuit of FIG. 9 is as follows. Suppose terminal1501 is connected to an AC source of 120 volts rms, that is swingingbetween −170 and +170 volts peak, and that terminal 1502 is connectedthrough a load to the neutral or return wire to the source. If theswitch is off, driver chip (1300) maintains a gate potential close tothe source potential. To turn the switch on, the driver chip applies apositive bias to the MOSFET gates of between 10 and 15 volts.

With the MOSFETs OFF, when terminal 1501 swings positive, intrinsicdiode 1102 becomes reverse biased and there is no current path totransistor (1200) or beyond. When terminal 1501 swings negative, diode1102 conducts and transfers the −ve voltage to the sources (s) of bothtransistors. However, intrinsic diode 1202 of transistor (1200) is thenreverse biased and there is no current path through to terminal 1502 orbeyond. Thus for both negative and positive voltage swings on the inputterminal relative to the output terminal, no current flows when theswitch is off. On the other hand, when gate driver chip (1300) makes thegates of transistors (1100,1200) positive with respect to their sources,both MOSFET channels (1101,1201) are in a low impedance state, allowingcurrent to flow from the input terminal to the output terminal with lowvoltage drop. The series loss resistance of the complete switch of FIG.9 is equal to the sum of the ON-resistances (RdsON) of the two MOSFETs.Each MOSFET needs to be able to withstand a voltage of twice the ACinput peak voltage in the OFF condition to survive the case of the inputand output being in anti-phase. Thus at least 340 volts is required,plus some margin. A suitable transistor is the Fairchild part numberFCH76N60N which has a 600 volt breakdown. Its RdsON is typically 28milliohms, giving a switch loss resistance of typically 56 milliohms.This gives just over 1 volt loss with 20 amps flowing, representing anefficiency of 99%. If desired, several such transistors can be used inparallel to reduce the loss resistance and allow for greater currentflow. For example, if each MOSFET were composed of three FCH76N60N:s inparallel, the voltage loss would be just over 1 volt with 60 ampsflowing.

Low pass filters (1001,1002) connecting the power sources to theswitches may comprise a series inductor and a parallel capacitor, theparallel capacitor being on the switch side. Low pass filter (1003) canalso comprise a series inductor and parallel capacitor, with the seriesinductor on the switch side and the capacitor on the output to theloads. This is because the switches alternately switch one source or theother into low pass filter (1003), and when the instantaneous sourcevoltages are slightly different, attempting to cause an instantaneousvoltage change on a capacitor would cause high current spikes, which isto be avoided. On the other hand, if there should be a very short periodof the order of 100 nS or so where both switches are off, any currentflow in the series inductor of filter (1003) attempts to continue toflow by causing a large voltage spike. Catching diodes would berecommended to limit the voltage spike. Greater circuit detail of theload-sharing arrangement of FIG. 4 is provided in FIG. 10 to elaborateon this issue.

FIG. 10 shows power coming from source 1 through a first LC filtercomprised of L1,C1 to a switch composed of MOSFETs 1100-1 and 1200-1,joined at their sources and gates. The gates are driven by opto-isolatedgate driver 1300-1, which may be a Fairchild part number FOD3180. Poweralso comes from source 2 through a second LC filter composed of L2,C2 toa second switch composed of MOSFETs 1100-2 and 1200-2 also with theirgates and sources commoned. The second switch gates are driven by asecond opto-isolated gate driver (1300-2). Gate drivers 1300-1 and1300-2 are powered by separate, mutually isolated DC supplies in the10-15 volt range. According as gate driver 1300-1 or 1300-2 iscontrolled to turn on its respective switch transistors, one or other ofthe selected sources is connected through 1200-1 or 1200-2 to the thirdlow pass filter composed of L3 and C3. According as gate driver 1300-1or 1300-2 is controlled to turn on its respective switch transistors,either source 1 or source 2 is selected to pass its voltage through tothe third low pass filter comprised of L3 and C3.

Now it is to be avoided that switch (1100-1,1200-1) be ON at the sametime as switch (1100-2,1200-2), and since it is difficult to guaranteeexact simultaneity, there is likely to be a time where both are OFF.During the “both OFF” period, L3 continues to attempt to maintaincurrent flow by making the voltage fly in such a direction as to keepthe current flowing. If the voltage is positive at the junction of1200-1 and 1200-2 and the current is flowing into L3 from that junction,the voltage will fly negative causing the intrinsic diodes of 1200-1 and1200-2 to conduct and pass the negative spike through to 1100-1 and1100-2. To avoid the voltage spike exceeding the breakdown voltage of1100-1 and 2, diode D2 of Spike Catcher circuit (1550) bypasses thecurrent into C5 causing it to charge negatively. Conversely, if thevoltage at L3 was negative and the current was out of L3 towards thejunction of 1200-1 and 1200-2, L3 will attempt to maintain the currentwhen both switches are off by making the voltage fly positive, riskingbreakdown of 1200-1 or 1200-2. To prevent this, Spike Catcher diode D1bypasses positive spike current into C4 causing it charge positively.The difference in voltage between C4 and C5 is constrained by zenerdiode Z1 (or some circuit equivalent to a zener diode) to be no morethan 400 volts, say, where that is less than the breakdown voltage ofthe MOSFETs.

Suppose the instantaneous load current is 84 amps, corresponding to thepeak of an AC current of 60 amps RMS, and suppose the period for whichboth transistors are off is 100 nS, and that that occurs upon each oftwo switch changes every 10 uS. Then that 84 amps is bypassed to eitherC4 or C5 for a total of 200 nS out of 10 uS, which is 2% of the time.The mean current is thus 2% of 84 amps or 1.68 amps at the peak of theAC cycle. Averaged over a half cycle gives the mean over a cycle of 1.68amps/π=535 mA into both of C4 and C5, Therefore Zener diode Z1 would bedissipating a power of the order of 535 mA times 400 v, or about watts.The AC actually flowing when the current is 60 amps rms is 120 volts×60amps, =7.2 Kw. The spike loss therefore is of the order of 1/36 of thepower, or around 2.7%. This reduces the efficiency of the circuit.Therefore alternative means to control spikes is sought that avoids thepower loss in spike catcher (1550).

To eliminate switching spikes, the four switch transistors may becontrolled in a deliberate sequence by using four independentlycontrolled gate drivers, as shown in FIG. 11. In FIG. 11, gate driver1300-1 controls only MOSFET 1100-1 while gate driver 1300-3 controls1200-1. Likewise, gate driver 1300-2 controls only MOSFET 1100-2 whilegate driver 1300-4 controls MOSFET 1200-2. Thus the timing of switchingeach MOSFET on or off is controllable independently.

Now suppose that the voltage at L3 is instantaneously positive and thatcurrent is being supplied through MOSFETs 1100-1 and 1200-1 from source1. To switch off switch (1100-1,1200-1), gate driver 1300-3 is firstcontrolled to switch off MOSFET 1200-1 while leaving MOSFET 1100-1 inthe ON state. The voltage at the junction of 1200-1 and 1200-2 will nowattempt to fly negative, causing the intrinsic diode of MOSFET 1200-1 toconduct and draw the current through the still ON 1100-1. Thus, despitehalf of switch (1100-2,1200-1) now being off, current still flows fromsource 1 to the load and there is no spike.

At the same instant, 1200-1 is turned off, 1100-2 may be turned on toprepare to take the current from source 2. If source 2 is momentarily ata slightly higher voltage than source 1, then it is the intrinsic diodeof 1200-2 that will catch the spike and maintain L3 current from source2 instead of source 1. In fact, the intrinsic diodes of OFF transistors1200-1 and 1200-2 form a diode-OR that passes whichever is the morepositive voltage, source 1 or source 2, to L3.

At this point transistor 1100-1 may be turned OFF as the current will bemaintained through the intrinsic diode of 1200-2 and ON transistor1100-2. Finally, transistor 1200-2 may be turned on to reduce itsvoltage drop to less than that of its intrinsic diode. Thus a successfultransition from switch (1100-1,12001) being ON with switch(1100-2,12002) OFF to the opposite state, without current or voltagespikes.

When the voltage from source 1 and/or source 2 to L3 is momentarily inthe negative part of the AC cycle, the switching sequence is as follows.

Suppose switch (1100-1,1200-1) is on and passing a negative voltage toL3 and passing a negative current from L3 to source 1. Switching thisswitch off would thus a cause a positive voltage spike at L3. To avoidthe spike, MOSFET 1100-1 is now turned OFF first while leaving 1200-1ON. The voltage at L3 attempts to spike positive, but is now caught bythe intrinsic diode of 1100-1, maintaining the current flow to source 1.At the same time, MOSFET 1200-2 may be turned ON to prepare to takecurrent from source 2. Current will actually then flow to whicheversource is at the more negative voltage, due to the diode-OR formed bythe intrinsic diodes of OFF transistors 1100-1 and 1100-2. MOSFET 1200-1may then be turned OFF and current will continue to flow through now ONtransistor 1200-2 and the intrinsic diode of 1100-2. Finally, MOSFET1100-2 is turned on to short out its intrinsic diode and reduce voltagedrop.

When the load is not resistive, the current through L3 will not beexactly in phase with the voltage. It is the direction of the currentthat preferably determines which of the above two sequences is used,rather than the sign of the voltage at L3. Thus the current directionthough L3 may be sensed with any suitable current sensor to control theselection of the switching sequence for that half cycle. If instead thesign of the voltage is used, the Spike Catcher (1550 of FIG. 10) willlikely still be needed to deal with reactive loads, but the loss ofpower in the Spike Catcher will now be proportional to the reactivepower, which is generally significantly smaller than the resistivepower, and so will not hurt efficiency so much. Since it is possiblethat load current could be zero, switching can also depend upon whetherthe current is of a sufficient magnitude to reliably use as a switchingsequence selection mechanism, with voltage sign being used by default.

FIG. 12 shows the switching waveforms of the four MOSFETs for thepositive and negative half cycles respectively.

The left half of the waveform diagram shows the switching sequenceduring a positive half cycle of the AC waveform (voltage or current, asdiscussed above).

A time t0, switch (1100-1,1200-1) is ON while switch (1100-2,1200-2) isOFF. At time t1. MOSFET 1200-1 starts to be turned off while almostsimultaneously at time t2, transistor 1100-2 starts to be turned on.When the current in L3 is positive, the current will attempt to continueto flow through the intrinsic diode of the now OFF transistor 1200-1. Assoon as 1100-2 is also ON, the current may also flow through theintrinsic diode of OFF transistor 1200-2 and the ON transistor 1100-2.The two sources are not connected together because both 1200-1 and1200-2 are off, and all that can happen is that their respectiveintrinsic diodes act to select the more positive of the source voltagesto pass to L3. After giving 1100-2 enough time to turn on, 1100-1 can beturned off at t3. Thus it is ensured that at least one source or theother is connected through to L3 via the intrinsic diode of 1200-1 or1200-2, thereby avoiding an inductive switching spike. When it iscertain that 1100-1 is OFF and 1100-2 is ON, 1200-2 is turned on at t4to short its intrinsic diode to reduce voltage loss. Switch(1100-1,1200-1) has thus been turned off and switch (1100-2,1200-2) hasbeen turned on, and will remain ON until t5.

At time t5, a reverse transition from switch 2 ON back switch 1 ON iscommenced. This is essentially the same sequence with the “−1” and “−2”transistors interchanged. After t8, switch 1 is back ON again and switch2 is OFF, and this state prevails until the second time labeled also t0to indicate a return to the beginning of the cycle.

The right half of the waveform diagrams of FIG. 12 show the switchingsequence during a negative half cycle (of current or voltage). Thesetimes are also labeled t0 to t8 but in the following description of thenegative half cycle switching sequence these times refer to thoseinstants in the right half of the waveform diagrams.

At the first time t0 of the negative half cycle, switch 1 is ON andswitch 2 is OFF. The changeover from switch 1 ON to switch 2 ON nowcommences at time t1 by turning off 1100-1. This is because a negativecurrent can now continue to flow through the intrinsic diode of 1100-1even though the transistor is OFF, thus averting a positive switchingspike. Almost simultaneously, at t2, 1200-2 is turned on. Now both1200-1 and 1200-2 are ON, but the two sources are not connected togetherbecause 1100-1 and 1100-2 are both off, and all that happens is that theintrinsic diodes of 1100-1 and 1100-2 act to select the more negativevoltage of source 1 and 2 to be passed through to L3. As both 1200-1 and1200-2 are ON at this point, a sufficient time must be allowed for1100-1 to be fully off before turning 1100-2 on at t3. After allowingsufficient time for 1100-2 to be fully on, 1200-1 may be turned off att4. This sequence ensures that there is always a path for negativecurrent to flow to either source 1 or 2 during the switching cycle,thereby avoiding high-voltage switching spikes.

The reverse transition from switch 2 ON and switch 1 OFF to the oppositestate begins at t5, where 1100-2 is first turned off as compared to1100-1 in the previous paragraph. Likewise the second action is to turn1200-1 on at t6 instead of 1200-2 on, as in the previous paragraph. thereverse transition is completed by turning 1100-1 on at t7 and 1200-2off at t8. Now switch 1 is back on while switch 2 is off, and thiscondition prevails until the second time t0 which is the start of aswitching cycle again.

FIG. 13 shows the circuit of a switching timing controller thatimplements the above switching sequence. Opto-isolared drivers 1300-1and 1300-3 have their LED input diodes connected back to back so thatwhen one is biassed ON the other must be biassed OFF. Opto-isolateddrivers 13002 and 1300-4 are connected likewise. High output currentbuffer type TC4468 supplies either 0 or a +5 v Vcc through resistors R1to R4 to these input LEDs. With a logic One (+5 v) at output pin 14 ofthe TC4468 and a logic Zero (0 volts) at pin 13, the input LED ofopto-driver 1300-1 will be turned on and MOSFET 1100-1 will be turned onwith MOSFET 1200-2 OFF. With a logic Zero at pin 14 and a logic One atpin 13 of the TC4468 MOSFET 1200-2 will be turned on and MOSFET 1100-1will be OFF. Output pins 11 and 12 of the TC4468 control MOSFETs 1100-2and 1200-1 likewise.

Microprocessor ATTINY461 is programmed to issue logic signals from itspins 11,12,13,14 respectively to control the corresponding output pinsof the TC4468. The sequence of logic signals issued can depend on eitherthe signs of the source voltages, which are conveyed to it throughSlicers 2001 and 2002 respectively, or the sign of the current measuredby current sensor (2000) conveyed to the microprocessor through slicer2003. If the sign of the current is positive, the sequence of logicsignals issued is as shown in the table below

1200- Pin 14 Pin 13 Pin 11 Pin 12 1100-1 1200-2 1100-2 1 1 0 0 1 ON OFFOFF ON 1 0 1 0 ON OFF ON OFF 0 1 1 0 OFF ON ON OFF 1 0 1 0 ON OFF ON OFFREPEAT FROM THE BEGINNING

The above table shows the previously discussed sequence required tochange from switch 1 on and switch 2 off to switch 2 on and switch 1 offwithout voltage spikes. This sequence is generated by executingmicroprocessor code to output bytes in quick succession to port PA bitsPA4,PA5,PA6,PA7.

Assume that the last byte output corresponds to the first line in theabove table, leaving switch 1 ON and switch 2 OFF. The processor thenwaits for the length of time for which SWITCH 1 is desired to be ON.After that time, the next two bytes are output, corresponding to lines 2and 3 of the above table, leaving switch 2 ON and switch 1 OFF. Themicroprocessor then waits for the length of time Switch 2 is desired tobe ON, and then outputs a further two bytes corresponding to the fourthand first line of the above table, returning to the state switch 1 ON,switch 2 Of F. This sequence then repeats until a logic change isdetected at PA1 from slicer 2003, indicating that the sign of thecurrent has changed. This causes an interrupt that branches the codesuch that it subsequently performs the sequence for switch changeintended for negative current half cycles shown in the table below.

1200- Pin 14 Pin 13 Pin 11 Pin 12 1100-1 1200-2 1100-2 1 1 0 0 1 ON OFFOFF ON 0 1 0 1 OFF ON OFF ON 0 1 1 0 OFF ON ON OFF 0 1 0 1 OFF ON OFF ONREPEAT FROM THE BEGINNING

The above description with the aid of FIG. 13 is intended to beexemplary and not an exhaustively researched and tested design. Inparticular, depending on the switch cycling frequency desired (i.e.lower or higher than 100 KHz) it may be that the ATTINY461microprocessor is not fast enough. In that case, a much faster processorsuch as an ARM, or else some digital hardware assistance would be inorder. The advantage of including a microprocessor at all is that it canhave a communications interface with other parts of the system, such asthe solar inverter/battery system, which can control the share of loadtaken from the solar system in dependence on the amount of solar energybeing received or on the state of charge of the battery. Other functionscan be implemented using the current and voltage sensors such asdetection of fault conditions such as excessive current, under voltageof one or other source, or source 1 and source 2 voltages having arelative phase error. The phase error can be measured by means of thevoltage zero crossings and transmitted back to an Inverter in order tosynchronize it.

Thus it has described above how a solid-state device can be constructedto share the burden of delivering current to an electrical load betweentwo independent power sources. The principle could be extended tosharing load current between any number of sources, using acorresponding number of AC switches such as in FIG. 9 and developedfurther in the above description. Of course for a split-phase 120/240volt system, it would be appropriate to use two instances of theinvention, one for each volt phase. For a three-phase system, threeinstances of the invention would be appropriate. In the split phase andthree-phase cases, multiple instances of the invention can sharecomponents such as the timing controller and be housed in a singleenclosure. This permits independent control of the power sharing foreach phase, as the load currents may be different for each phase. Alsoin mutli-phase systems, there may be one inverter per phase, amulti-phase Inverter, and such inverters may be powered from the same ordifferent batteries or solar arrays, also making the optimum powersharing to be achieved different for each phase.

For completeness, another possible circuit for an AC switch is shown inFIG. 14. A diode bridge (3002,3003,3004,3005) ensures that, whatever thepolarity of the source, a positive voltage is applied to the drainterminal D of N-type MOSFET (3000) and a negative voltage is applied tothe source terminal S. When MOSFET (3000) is OFF, no current flows inthe bridge and therefore no current flows in the source. When MOSFET(3000) is ON, the bridge output is short circuited and the bridgepresents a low AC impedance from the source to the load. The voltagedrop of the switch of FIG. 14 is two diode drops plus the drop of MOSFET(3000). This may be larger than the voltage drop of the switch of FIG.9, but when this drop is small compared to the operating voltage, whichin some applications may be higher than volts, FIG. 14 can be a lowercost solution.

Higher voltage switches can be built by combining semiconductors inseries, in either FIG. 9 or FIG. 14. Thus the invention could beextended to utility scale transmission voltages such as 13,200 volts orhigher, providing a “smart grid” fine control over load sharing betweentwo or more generating sources.

When one of the sources, say source 2, is a DC to AC Inverter, it islikely to be of a switching type that already has a low-pass filter atits output. It is not necessary to have both an inverter output filterand the low pass filter (1002) of FIG. 4. A single filter can be used,that has a cut off frequency related to the lower of the inverterswitching frequency and the switching frequency used in the invention.Moreover, it may be possible to absorb switch (1010-2) of FIG. 4 intosuch an inverter, if the inverter is not powering anything else.

FIG. 16 shows an arrangement where one of the AC switches and low-passfilter (1001 or 1002) of FIG. 4 are absorbed into the DC to AC inverter.Filter (1001 or 1002) would have normally existed at the output of aninverter to smooth any switching steps and maintain a good sine waveapproximation at the power line frequency. This filter is now combinedwith filter (1003), as it suffices that the inverter output pass throughonly one filter on the way to the load.

An inverter such as described in the herein-incorporated '822 patent mayutilize a switched approximation to a sine-wave, such as deltamodulation, delta-sigma modulation or pulse-width modulation. Theswitched approximation is produced by an H-bridge composed oftransistors Q1,Q2,Q3,Q4 in FIG. 16. To obtain a sine wave at the powerline frequency, Q1 is turned on with Q2 off when a negative half cycleis to be produced, connecting the positive of the 170 v DC source toneutral. The Live output is produced by switching Q3 and Q4 alternatelyto produce −170 v or 0 at the output to L3. The minimum time spent ateither 0 of −170 v is limited to being not less than a minimum value,such as 1 uS, while controlling the proportion of time that the outputis switched to −170 should equal the ratio that the instantaneous valueof the desired output 120 v rms sine wave bears to the 170 volt peak.The positive half cycle is produced by turning on Q2 with Q1 off,connecting the DC −ve to neutral. Q3 and Q4 then switch +170 v DC orzero through to L3 for proportions of the time required to approximate asine wave, with the same constraint of a minimum dwell time at one orthe other level. Mathematically, the volt-time integral over the periodthat the 170 v DC supply is switched through to L3 should equal thevolt-time integral of the desired 120 v rms AC sine wave. Such an ON-OFFsequence can be precomputed and stored in memory. A good method ofprecomputing the sequence is to use a multi-state Viterbi look-aheadmethod of generating sigma-delta modulation, as described in, forexample, “Efficient trellis-type sigma-delta modulator”, P Harpe, D.Reefman and E. Jannsen, Proceedings of the Audio Engineering SocietyConvention, 2003.

If both Q3 and Q4 are switched OFF, when there is current in L3, thevoltage spike is caught either by the intrinsic diode of Q3 or that ofQ4. The stored energy in L3 is actually then returned to the DC source.Therefore no separate spike catching circuit is required and there is noloss of efficiency entailed. When Q3 and Q4 are both OFF moreover, ACswitch (1010-1) may be turned ON to maintain load current from source 2,e.g. from the electricity grid. If the peak voltage of the electricitygrid is greater than the 170 v DC supply voltage, the intrinsic diodesof Q3 and Q4 form a full-wave rectifier feeding power backwards to theDC source. This can be used to keep a battery float charged for use inan emergency, but if this not desired, a blocking diode can be insertedin one or other DC supply lead to prevent reverse current flow. Adecoupling capacitor is then required on the H-bridge side of theblocking diode to recover reverse current flow energy.

Thus using the invention in the form of FIG. 16 to share power between afirst source such as a solar inverter and a second source such as theelectricity grid comprises alternating between periods during which Q3and Q4 are OFF and AC switch (1010-1) is ON, and periods during which ACswitch (1010-1) is OFF and Q3 and Q4 go on and off alternately togenerate an instantaneous approximation to the desired sine wave powerfrequency voltage. In the configuration of FIG. 16, the switching of theinverter and the switching of the AC switch therefore have to becoordinated to achieve the goals of load sharing in the desired ratio byutility and inverter respectively, maintaining a good sine wave to theload, and maintaining a sine-wave current draw from the utility, withlow harmonic content and without exporting high frequency interference.

For completeness it is considered whether any other coupling deviceother than the arrangement of semiconductor switches disclosed abovecould be contemplated for coupling a first power source and a secondpower source to a load to thereby share the load current without thesources being coupled together. FIG. 15 shows an arrangement of twofixed coils at right angles and a rotatable coil. From other fields ofelectronics, such an arrangement is known as a goniometer. The rotatingcoil 4003 can be rotated to be in the plane of fixed coil 4001, and thusstrongly coupled to it and not coupled to coil 4002, or can be rotatedto be in the plane of coil and not coupled to coil 4001. In between, itis coupled to both in the ratio Cosine(θ) to one and Sine(θ) to theother.

A goniometer is generally used at radio frequencies as part of adirection finding apparatus. To use a goniometer at power linefrequencies, a high permeability core would be needed to increase coilinductance values. For example, a laminated iron core can be used, andthe whole arrangement enclosed in a laminated iron outer pot to completethe magnetic circuit. A motor Stator/Rotor configuration could also beused. Thus an electromechanical arrangement can serve as a couplingdevice to couple two sources to the same load in variable proportionswithout coupling the sources to each other.

The voltage picked up by the rotatable coil is V3=V1 cos(θ)+V2 sin(θ),where θ is the angle between the rotating coil 4003 and fixed coil 4001,and V1 is the voltage of source 1 and V2 is the voltage of source 2. IfV1 and V2 are in phase, V3 is not independent of angle because theweighting factors do not add to unity, as did the weighting factors “a”and “b” in the foregoing arrangements. In order for the output voltageto remain between V1 and V2 as the rotating coil is moved, V1 and V2 canbe deliberately maintained 90 degrees out of phase. Then the magnitudeof V3 is given by

√(V1 cos(θ))²+(V2 sin(θ))²

which is constant when V1=V2 because (V1 cos(θ))²+(V2 sin(θ))²=1, andvaries between V1 and V2 as θ is rotated otherwise.

When the two source voltages are 90 degrees out of phase, a rotatingmagnetic field if produced that will attempt to rotate the iron core ifeddy currents are induced. If the iron core is left free to rotate, itwill rotate at the power line frequency (3000 RPM for 60 Hz) and no eddycurrent losses will then be incurred.

There is also a strong torque applied to the rotatable coil proportionalto the load current. This requires that the rotating coil be firmlyanchored when it is placed at the desired angle.

The arrangement of FIG. 15 maintains strict isolation between thesources, the current from source 1 being equal to the load current timescos(θ) and independent of V2 while the current from source 2 is the loadcurrent times sin(θ) and independent of V1. The device of FIG. 15 is nothowever necessarily more efficient than the switching method describedheretofore, and may be much larger and more expensive for the same powerrating. This is likely also to be the case if an attempt were made torealize FIG. 6 using switched-tap transformers, however, it might be acompetitive realization for very high power utility applications.

Thus a useful method and apparatus has been described for sharing theburden of providing an electrical current to a load. The method can beadapted to work for DC power sources as well as AC power sources. Aperson skilled in the art may make many adaptations of the circuitsdisclosed without departing from the spirit of the invention which iscaptured in the attached claims.

I claim:
 1. An apparatus for adaptively sharing an electrical loadbetween two independent, electrical power sources, comprising:— Avariable coupling device to couple a first electrical power source tosaid load to supply a first proportion of the load current, and tocouple a second electrical power source to the same said load to supplya second proportion of said load current, the coupling devicemaintaining isolation between the first and second power sources suchthat no power is transferred from the first power source to the secondpower source or vice versa.
 2. The apparatus of claim 1 in which saidvariable device further comprises:— A first low pass filter forconnecting said first electrical power source to a first input of aselection switch; a second low pass filter for connecting said secondelectrical power source to a second input of said selection switch;output terminals for connection to said load, the output terminals beingconnected to the output of said selection switch through a thirdlow-pass filter, and a switch controller for causing said selectionswitch to select said first and second power source alternately at ahigh rate with respect to the cut-off frequency of said first, secondand third low pass filters, wherein the proportion of time a source isselected determines the proportion of the load current supplied by thatsource.
 3. The arrangement of claim 1 configured for said first andsecond electrical power sources being AC sources approximatelysynchronized in voltage and phase.
 4. The apparatus of claim 1configured for said first and second electrical power sources to be anelectrical utility grid supply and a DC to AC inverter, respectively. 5.An apparatus for adaptively sharing an electrical load between twoindependent, alternating current electrical power sources, comprising:—A first low pass filter for connecting said first electrical powersource through a first electrically controlled semiconductor AC switchto the input of a third low pass filter; a second low pass filter forconnecting said second electrical power source through a secondelectrically controlled semiconductor AC switch to the same input ofsaid third low pass filter; output terminals for connection to saidload, the output terminals being connected to the output of said thirdlow-pass filter, and a switch controller for controlling said first andsecond AC switches to pass current from said first and second powersource alternately at a high rate with respect to the cut-off frequencyof said first, second and third low pass filters, wherein the proportionof time a source is selected determines the proportion of the loadcurrent supplied by that source.
 6. The apparatus of claim 5 in whichsaid first and second semiconductor AC switches each comprise: an inputterminal connected to the drain of a first MOSFET; an output terminalconnected to the drain of a second MOSFET; a floating DC voltage sourceto provide a gate drive potential for said first and second MOSFETs, onepolarity output of said floating voltage source being connected to thesource terminals of both said first and second MOSFET and the oppositepolarity of said floating voltage source being connected to the powersupply input for a gate driver circuit, the gate driver circuitsupplying gate drive outputs to the gate terminals of said first andsecond MOSFETs and providing electrically isolated control inputs forreceiving control signals from said switching controller.
 7. Theapparatus of claim 5 in which said first and second semiconductor ACswitches each comprise: an input terminal connected to the drain of afirst MOSFET having an intrinsic drain-source diode; an output terminalconnected to the drain of a second MOSFET having an intrinsicdrain-source diode; a floating DC voltage source to provide a gate drivepotential for said first and second MOSFETs, one polarity output of saidfloating voltage source being connected to the source terminals of bothsaid first and second MOSFETs and the opposite polarity of said floatingvoltage source being connected to the power supply input for a gatedriver circuit, the gate driver circuit supplying gate drive outputs tothe gate terminals of said first and second MOSFETs and providingelectrically isolated control inputs for receiving control signals fromsaid switching controller, wherein said switching controller controlssaid gate drivers of the first and second MOSFETs of each of said firstand second AC switches to turn ON and OFF in such a sequence that at notime are both MOSFETs of both switches ON at the same time and such thatthe current flowing between the output terminal of the first or secondswitch and said third low pass filter always has a path through one orother of said MOSFETs or said intrinsic darin-source diodes.
 8. A methodfor adaptively sharing an electrical load between two independent,electrical power sources, comprising: Determining a first fraction ofthe load current to be provided by a first of said two independentelectrical power sources; determining the remaining fraction of the loadcurrent to be provided by the second of said two independent electricalpower sources Transforming the voltage output of said first electricalpower source to provide said first fraction of the desired load voltage;transforming the voltage output of said second electrical power sourceto provide said second fraction of the desired load voltage; connectingthe transformed voltage outputs of said first and second power sourceseffectively in series such that said first and second fractions of saiddesired load voltage add to produce said desired load voltage.
 9. Themethod of claim 8 in which said determined first fraction of the loadcurrent is provided by a solar energy source, and the correspondingfirst fraction of the power is controlled to match either the average orthe instantaneous solar energy received.
 10. The method of claim 8 inwhich transforming the voltage output of said first electrical powersource means controlling a first switch to select the output of saidfirst electrical power source to be connected for said first fraction ofthe time to said electrical load through a low pass filter andtransforming the voltage output of said second electrical power sourcemeans controlling a second switch to select the output of said secondelectrical power source to be connected for said second fraction of thetime to said electrical load through said low pass filter, said firstand second switches never selecting to connect respective sourcesthrough said low pass filter to said load at the same time.
 11. Themethod of claim 8 in which transforming the voltage outputs of saidfirst and second electrical power sources means controlling a firstswitch and a second switch alternately to select the output of saidfirst electrical power source to be connected for said first fraction ofthe time to said electrical load through a low pass filter and to selectthe voltage output of said second electrical power source to beconnected for said second fraction of the time to said electrical loadthrough said low pass filter, the alternate selection occurring at sucha frequency that said low pass filter substantially prevents anyunwanted voltage ripple related to said frequency from appearing at theload.
 12. The method of claim 8 in which transforming the voltageoutputs of said first and second electrical power sources meanscontrolling a first switch and a second switch alternately to select theoutput of said first electrical power source to be connected for saidfirst fraction of the time to said electrical load through a low passfilter and to select the voltage output of said second electrical powersource to be connected for said second fraction of the time to saidelectrical load through said low pass filter, the alternate selectionoccurring at such a rate that said low pass filter substantiallyprevents any unwanted voltage ripple at said rate from appearing at theload, and using further low pass filters between said first power sourceand said first switch and between said second power source and saidsecond switch such that said first and second power sources experience amean current load substantially free from ripple at said alternatingswitch selection rate.